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Full text of "tektronix :: concepts :: 062-1056-00 Oscilloscope Trigger Circuits Feb69"

Circuit 
Concepts 





OSCILLOSCOPE 
TRIGGER 
CIRCUITS 

BY 
HAROLD MAGEE 




CIRCUIT CONCEPTS 



FIRST EDITION FEBRUARY 1969 
062-1056-00 
PRICE $1.00 



©TEKTRONIX, INC.; 1969 
BEAVERTON, OREGON 97005 
ALL RIGHTS RESERVED 



CONTENTS 



1 INTRODUCTION 1 

2 TRIGGER CIRCUITS 5 

3 INPUT TRIGGERING SIGNALS 23 

4 PULSE GENERATORS 43 

5 DELAYING AND DELAYED SWEEPS 57 

6 TRIGGERED DELAYED SWEEPS 73 
INDEX 81 

CIRCUIT CONCEPTS BY INSTRUMENT 83 













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HORIZONTAL 








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INTRODUCTION 



In a study of conventional oscilloscopes, the various 
circuits of the instrument will fall into a few 
general groups. These may be listed as: cathode-ray 
tube, power supply, vertical amplifier, horizontal 
amplifier, sweep generator, and the trigger circuit. 

Each part performs an important function and, in part, 
determines the final performance of the instrument. 

THE The cathode-ray tube displays light on a two- 

CATHODE-RAY dimensional phosphor screen which conveys 
TUBE intelligence in the form of graphs, alphanumerics 
or picture images. Graphical presentations offer 
an analytical approach in that actual measurements 
can be taken with a graticule along the screen's 
"x" and "y" axis. 

The CRT electron gun is sealed inside an envelope 
CRT and a vacuum is created to minimize collisions 

electron between free gas particles and the electron beam. 
9 un High-voltage power supplies connected to the CRT 

create controllable electrostatic fields which 
accelerate free electrons from a heated cathode 
to form an electron beam. The beam of electrons 
then transit an electron lens which converges or 
focuses the beam on a phosphor screen. When the 
high-velocity electrons collide with phosphor atoms 
at the focal point, photons of light are emitted 
towards the viewer. 

The positioning of the point light source on the 
screen is accomplished by varying the electrostatic 
field between a set of "x" and "y" deflection plates. 

electrons This permits the electron beam to create a display 
anywhere within the viewing screen area. Since 
electrons have an extremely small mass, they can be 
deflected or scanned over the entire screen area 
millions of times per second. This permits the 

real time viewer to observe changing phenomena in real time. 



The CRT must not present an undue loading on either 
the vertical or horizontal amplifier or require a 
greater dynamic-deflection voltage range than the 
amplifiers can supply. In practice, the CRT and 
deflection amplifiers must be designed together for 
maximum performance and efficiency. 

VERTICAL The vertical amplifier of an oscilloscope determines 
AMPLIFIER the useful bandwidth and gain of the instrument. The 
vertical amplifier may take four general forms: a 
fixed vertical that cannot be changed, a complete 
vertical in a plug-in form that may be changed, or 
a fixed main amplifier that is preceded by a plug-in 
preamplifier that can be changed to have different 
characteristics. An additional type of instrument 
takes the drive directly to the CRT plates without 
passing through any type of amplifier. 

The general purpose oscilloscope is intended to 
provide a faithful display of an input pulse. For 
meaningful results, displayed waveforms must contain 
few aberrations and these must be but a few percent 
of the total waveform amplitude. 

The following characteristics are important in 
describing the performance of a vertical amplifier. 

bandwidth Bandwidth (BW) describes the gain versus 

frequency limits between an upper and lower 
frequency. These frequencies are the points 
where voltage gain drops to 70.7% of maximum 
(3 dB down) . To have a lower frequency -3 dB 
point, the scope must be AC coupled. The low- 
frequency response is that of a high-pass RC 
filter. The upper-frequency rolloff follows 
a curve that usually approximates the gaussian 
error curve. Amplifier low-frequency response 
extends to DC. Therefore, the upper -frequency 
-3 dB point describes the bandwidth for a DC- 
coupled instrument. 

risetime Eisetime (Tr) indicates the transient response 
characteristics of a vertical amplifier. One 
measures risetime along the leading edge of a 
displayed voltage step. This specifies 
transition time from 10% to 90% of maximum 
voltage. Bandwidth of a general-purpose 



instrument can usually be related to the 
risetime by the equation BW • T^ = .35. 

deflection Defleation factor describes the voltage required 
factor at the input to deflect the trace one CRT 

division and is expressed as V/div. 

THE The horizontal amplifier converts the sweep-generator 

HORIZONTAL time-base ramp to a driving signal for the horizontal 
AMPLIFIER deflection plates of the CRT. In the process it 

must transform the single-ended input signal to an 
amplified push-pull signal with minimum departure 
from linearity and at the required deflection factor. 
X-Y In those oscilloscopes offering (X-Y) presentations 

presentation at the full bandwidth of the instrument, the 
horizontal amplifier must exhibit a frequency 
response comparable to that of the vertical amplifier. 
Y-T displays In conventional Y-T displays, it provides direct- 
current (DC) level controls, through which the overall 
gain may be multiplied by factors of one or more, 
providing a sweep speed increase by the same factor. 



SWEEP 
GENERATOR 



time 

measuri ng 
reference 



The sweep generator produces a sawtooth waveform 
that is processed by the horizontal amplifier to 
deflect the CRT electron beam across the face of 
the CRT. The major function of the sweep generator 
is to produce a sawtooth waveform with the proper 
rate-of— rise amplitude and linearity that will 
provide a suitable time measuring reference for the 
cathode—ray oscilloscope. 



TRIGGER The input signal may have a wide variety of shapes 
CIRCUIT and amplitudes, many of which are unsuitable as 

sweep initiating triggers. For this reason these 
signals are first applied to a trigger circuit where 
they are converted to pulses of uniform amplitude 
and shape. The addition of a trigger circuit makes 
it possible to start the sweep with a pulse that 
has a constant size, eliminating variations of the 
changing sweep-circuit operation caused by changing input 
input signals. This extra circuit allows the operator to 
signals use either slope of the waveform to start the sweep, 
select any voltage level on the rising or falling 
slope of the waveform, and, in some instances, 
filter out selected frequencies of the input signal 
with greater ease and repeatability. 



TO VE 
AMPL 


TTICAL 
FIER 














TRIGGER 


INPUT 
COUPLING 
CIRCUITRY 




INPUT 
AMPLIFIER 




PULSE 
GENERATOR 


CIRC 


JITRY 









TO SWEEP 
GENERATOR 



EXTERNAL , 

NPUT 



Fig. 2-1. The trigger circuit in the 
oscilloscope. 



TRIGGER CIRCUITS 



block The general block diagram (see Fig. 2-1) of an 
diagram oscilloscope triggering circuit can be separated into 
four basic parts: vertical-amplifier trigger takeoff 
circuitry, input-coupling circuitry, input amplifier 
and pulse generator. The complexity of each section 
is determined by the triggering requirements of the 
oscilloscope. These requirements are basically 
established by the vertical amplifier characteristics 
and the sweep speeds available from the horizontal 
amplifier. 

trigger Any oscilloscope that can be triggered internally 
takeoff from the viewed signal must have some type of 
circuitry coupling to transmit a sample of the vertical signal 
to the input of the trigger circuit. This circuitry 
may take several forms depending on performance 
requirements of the oscilloscope. It may vary from 
a very simple divider to a complex push-pull amplifier. 

In each case the general requirements of the circuit 
are the same. The takeoff circuitry must act as a 
requirement buffer, to keep the trigger circuitry from changing 
the operation of the vertical amplifier and yet pass 
the vertical-amplifier signal to the trigger circuit, 
without undue amplitude or frequency distortion for 
the operating ranges of the oscilloscope trigger. 
The trigger takeoff circuitry may also have to change 
voltage and gain levels between the vertical amplifier 
and the trigger. In addition, the trigger circuitry 
must not add noise or other interfering signals to 
the triggering signal at an amplitude sufficient to 
cause erratic triggering. Stable operation of the 
trigger circuit, when triggering internally, is in 
part, dependent on the stability of the takeoff 
circuit. 



no i se 



B+ 
+250V 



TRIGGER TAKEOFF 



30 k 




470 k 



330k 




VERTICAL 


CRT 


AMPLIFIER 


DEFLECTION 


OUTPUT 


PLATE 



Fig. 2-2. Simple trigger-takeoff circuit. 



VERTICAL 

OUTPUT 

AMPLIFIER 



3.9pF 



, „ INTERNAL 
250k^" r TR | GGER 

DC LEVEL 



+300V -100V 



CRT 

DEFLECTION 

PLATE 




100V 



TRIGGER 
OUTPUT 



Fig. 2-3. Trigger-takeoff circuit for 

oscilloscope without delay line. 



Fig. 2-2 illustrates one of the simple types of 
trigger trigger takeoff. This circuit takes the output 
takeoff directly from one CRT plate in a low frequency 
oscilloscope. There is no delay line in the 
instrument, so it can be triggered from the vertical 
output stage with no need for additional gain. The 
divider network is not compensated for high frequencies 
and the loading on one side of the vertical amplifier 
is not a problem at low frequencies. The network 
provides an attenuated output near zero volts with 
the trace centered on the cathode-ray tube. 

low- Fig. 2-3 is a trigger takeoff circuit from an 

frequency oscilloscope without a delay line. The takeoff 
point is at the CRT deflection plate so no 
amplification is needed. The divider has a 3.9 pF 
capacitor to compensate it for high frequencies and 
the resistive load is less than the low-frequency 
circuit in Fig. 2-2. An adjustment is added to set 
the cathode-follower output at zero volts with the 
trace centered. The cathode follower reduces the 
loading on the vertical amplifier and provides a 
low- impedance output to the trigger circuit to retain 

high- the high-frequency component of the triggering signal. 

frequency In this circuit, both alternating current (AC) and 
DC triggerlng-signal amplitudes receive the same 
attenuation. 




TRIGGER 
OUTPUT 



1 >• TR I GGER 

OUTPUT 



Fig. 2-4. High frequency pickoff circuit 
with a delay line. 



Fig. 2-4 is the schematic of a pickoff circuit from 
a high-frequency oscilloscope with a delay line. 
There is a low impedence-takeof f point from each 
side of the vertical amplifier followed by cathode 
followers to minimize loading and provide an equal 
load to each side of the amplifier. 

Fig. 2-4 The oscilloscope referred to in Fig. 2-4 has a wide- 
frequency response and the takeoff system must pass 
these frequencies to the trigger circuit, when the 
instrument is operated in either the triggered or 
synchronized mode. VI and V2 provide added 
amplification for the signal. 

The plate load for V2 is an inductive resistance to 
provide good high-frequency response. An advantage 
of push-pull, trigger-takeoff operation is that it 
provides common-mode signal rejection so that 
variations due to DC level changes and power supply 
variations are not transmitted to the trigger circuit. 
A cathode follower is used after the amplifier stage 
to reduce loading, produce a low-impedence output, 
and retain the wide bandpass necessary for the 
trigger-input signal. A second cathode follower 
provides an AC-coupled, vertical-signal output at the 
front panel of the oscilloscope and is a takeoff 
point for a second trigger circuit, if the instrument 
has a delayed sweep. 

delay line The function of the delay line is to provide enough 
time delay for the vertical signal so that it will 
not reach the vertical-deflection plates of the CRT 
before the trigger and sweep circuits have been 
activated by the input signal. If there is not 
enough delay in the vertical amplifier it will not be 
possible to see the first part of a fast signal on 
the CRT. 

In order for the delay to be effective, the trigger 
signal must be taken from some point that precedes 
the vertical delay. 




Fig. 2-5. Complex- takeoff circuit with 
push-pull output signal. 



11 



Fig. 2-5 illustrates a takeoff circuit that is more 
cascode complex. This circuit produces a push-pull output 
amplifier signal to the trigger-input amplifier. Transistors 
Q3 and Q103 operate as emitter followers for the 
vertical amplifier and as the lower half of cascode 
amplifiers for the takeoff circuit. The upper half 
of each cascode amplifier drives an emitter follower 
which sends a signal to the trigger-input amplifier. 
The output level of the cascode amplifier is sensed 
and amplified by regulator Q83/Q84 and applied to the 
cascode amplifier input to cancel out the change, 
which returns the circuit's average-output level to 
the same point. The voltage variation then appears 
across Q74 and Q17A. This keeps common-mode signals 
from upsetting the levels, at the trigger circuit, 
"plug- in" when different plug-ins are inserted in the 
levels oscilloscope, or when the common mode signal-level 
shifts. 

single Some multi-channel instruments have the option of 
channel, triggering from one channel, this enables the operator 
trigger to start the sweep each time with the same signal and 
takeoff makes accurate time measurements between signals on 
the other channels possible in either alternate or 
chopped modes of operation. Plug-ins that have this 
capability may have an output on the front panel that 
can be connected to the external-trigger input on the 
main frame to provide this mode of operation in 
^ instruments lacking an internal connection. 



12 



amp I i f i er 
bandpass 



Since this trigger takeoff point is closer to the 
input of the oscilloscope, it is necessary to add more 
amplification, than when the pickoff is from a higher 
signal level. The added amplification requirements 
may result in a more restricted bandpass from this 
amplifier. A typical amplifier bandpass is shown in 
Fig. 2-6. 




2MHz 4MHz 5.5MHz 
FREQUENCY *■ 



Fig. 2-6. Typical amplifier bandpass. 



The Fig. 2-6 also shows that the amplifier has a 
dynamic range limitation. This requires that high- 
frequency triggering and extremely low-frequency or 
DC DC triggering should be done from the main amplifier 

triggering triggering point. Some single-channel amplifiers, 

however, do cover the entire triggering range of the 
oscilloscope. In most cases the positioning control 
for the individual channel comes after the pickoff 
point in the amplifier, so the trigger-output level 
from a single channel is not changed by positioning. 
As a result of this, DC triggering does not appear to 
work normally. 



13 



norma I 
triggering 



DC 

triggering 
from 
Channel 1 



In the internal-normal, DC-coupled mode, the 
triggering point or trace-starting point is related 
to a specific level on the CRT screen. As the 
waveform is positioned up and down on the screen, 
the triggering point changes on the waveform, always 
being related to the same vertical level on the 
screen. If the waveform is placed so that it is 
all above or all below this point, triggering will 
cease. An instrument that is DC triggered from 
Channel 1 will not act this way, because the level 
information is added to the vertical signal after 
the Channel 1-pickoff point and position does not 
affect the triggering point. See Fig. 2-7. 



POS I T I ON 
CONTROL 







VW. 












OF 

INPUT 




PREAMPLIFIER 




MAIN 
AMPLIFIER 




































TRIGGER 
PICKOFF 


























CHANNEL 1 
AMPLIFIER 




TRIGGER 
NPUT 












P 


_UG- 1 N 





Fig. 2-7. Triggering Position/Control. 



1A 



TO VERTICAL 
AMPL I F I ER 



TO CHANNEL 1 
INPUT 

ATTENUATOR 

^n — ^ 




Fig. 2-8. Individual channel trigger-pickoff 
circuit . 



15 



amplifier Fig. 2-8 is a simple type of individual-channel, 
gain trigger pickoff. The amplifier is composed of two 
operational-amplifier stages. The gain of the 

amplifier is closely approximated by _£. In the first 



stage the gain is 2 ' 7K SI - 2.3. The gain of the 

second stage is frequency dependent, as the 25 yF 
capacitor-reactance adds to the 510 fi R-^-resistance. 
At a frequency where the reactance of the capacitor 
is low enough to be neglected, the gain of the 

3900 ( R /) 
amplifier is — ) ' ~ 7.6. Multiplying this 

gain, by the first-stage gain, results in a gain of 
about 17.5. Reactance of the 25 uF capacitor causes 
the response of the amplifier to rolloff. At a 
frequency of 10 Hz the signal is down 3 dB. When 
used with a "plug-in" with a basic sensitivity of 
50 mV per centimeter, it has an output in excess of 
0.5 volts per centimeter. The amplifier is AC 
coupled as discussed above, which limits its low 
frequency response but has the advantage of being 
less susceptible to overloading due to DC offsets at 
the input of the channel. 

-f- r j aaer Early oscilloscopes used only AC coupling between 
co I the trigger-pickoff circuit and the trigger-input 
cjr cuitrv amplifier. This type of coupling was satisfactory 

in many cases, but as sweep generators became capable 
of generating slower sweeps, it was necessary to add 
a coupling circuit that would respond to lower 
frequencies. The addition of DC coupled-vertical 
amplifiers also made it convenient to be able to 
trigger at a set level on the screen. 

Other refinements were added. AC low frequency 
reject made it possible to trigger on high frequency 
signals in the presence of low frequency signals; 
it also was a triggering aid when two input channels, 
switching alternately, were added to the input of the 
vertical amplifier. Some oscilloscopes also have a 
high frequency reject position. This allows the 
operator to reject high frequency signals or 
transients and start the sweep on lower frequency 
signals only. 



16 



EXTERNAL | 
IliPUT 



TRIGGER 
TAKEOFF 



CI 
.01 

Trlh- 



C2 
.0022yF 



> R2 
PIN- 



TRIGGER 
CIRCUIT 



Fig. 2-9. An AC coupling circuit. 




I i 



0Hz 100Hz 



100kHz 1MHz 7MHz 10MHz 



Fig. 2-10. AC-coupled trigger response. 



EXTERNAL 
INPUT 



TRIGGER 
TAKEOFF 



.0022 



100pF 



:1M ?100k 



TRIGGER 
CIRCUITS 



Fig. 2-11. Low frequency reject circuit. 



17 



A typical AC coupling circuit is shown in Fig. 2-9. 

resistance- There are two series resistance-capacitance (RC) 
capacitance circuits, the one composed of Cl/Rl is only used to 
circuits keep C2 from charging to +300 volts and discharging 

through any circuit connected to the External trigger 
input when the trigger-source switch is switched to 
the External position. C2/R2 are the primary coupling 
and limit the low frequency operation of the circuit. 
High frequency response is limited primarily by stray 
circuit capacities. A typical circuit may have 
-3 dB response points at 100 Hz and 7 MHz as shown 
in Fig. 2-10; the actual rolloff frequencies will 
vary with instrument type. The low frequency -3 dB 
point of the amplifier is equal to the frequency 
that causes the capacitive reactance (X G ) of C2 to 
equal the resistance of R2. The actual frequency may 
vary somewhat as the two components are not precision 
parts. This does not mean that triggering stops at 
these frequencies, but that increased signal 
amplitude will be required below these frequencies 
for stable triggering. The RC rolloff at the low 
frequency end of the bandpass is 6 dB per octave, 
which is the same as 20 dB per decade. The high 
frequency end of the bandpass is determined by many 
factors and is not an RC rolloff, but more closely 
approaches a Gaussian rolloff. The factors which 
regulate the high frequency rolloff are wiring 
capacitances, lead inductances and the high frequency 
characteristics of the amplifying device. 

An example of a low frequency-reject circuit is shown 
in Fig. 2-11. This may also be called AC Fast on 
some instruments. 

low In this circuit a 100 pF capacitor has been added in 

frequency- series with the input and the 1 M input resistor of 
reject the trigger circuit has been paralleled with a 100 K 
resistor. This time constant change causes the lower 
3 dB frequency to be shifted from 100 to 10 kHz. 
This attenuates low frequency-sine waves and 
differentiates low frequency-square waves or pulses. 
Because of this it is possible to use a dual trace 
instrument more easily in the alternate mode with the 
trigger signal originating from the main-vertical 
alternate amplifier. The input-time constant is selected so 
mode that the square wave caused by switching from channel 
to channel is differentiated, but recovers to a 
quiescent level during the sweep holdof f period 



18 



AC MODE 

TRIGGER INPUT 



LF REJECT MODE 
TRIGGER INPUT 



r 



r 



k 



r 



SWEEP 
WAVEFORM 




HOLDOFF 



Fig. 2-12. Waveform produced by selecting 
alternate mode triggering. 



See Fig. 2-12. During this time the trigger circuit 
may produce several outputs, however the signal 
produced at the end of holdoff is caused by a signal 
applied to either of the channels and is not a result 
of alternate switching waveforms. 




lOMrlz 



Fig. 2-13. Low frequency reject-trigger 
response . 



Another aid to triggering is to position the traces 
as closely together as possible for the measurement 
being made. This reduces the amplitude of channel- 
switching transition fed into the trigger circuit. 

Fig. 2-13 shows a typical response for the trigger- 
circuit input in the low frequency-reject mode. 



19 



Since the range of AC triggering drops off rapidly 
DC coupling at frequencies below 100 Hz, it may be difficult to 

trigger on slowly changing waveforms with AC coupling. 
The addition of DC coupling extends the low frequency 
triggering capabilities to the slowest waveforms. 
A typical DC coupling circuit is shown in Fig. 2-14. 




DC LEVEL 



-150V 



Fig. 2-14. DC-coupling circuit. 



The output of the trigger takeoff circuit has a level 
of about +320 V, when the cathode-ray tube trace is 
centered. This level must be reduced so that the 
input to the trigger circuit is zero volts with the 
trace centered. This makes zero volts-input from 
the internal circuit correspond to zero volts-input 
internal from the external input. In Fig. 2-14 a resistive 
trigger DC divider is used to accomplish this. An adjustment, 
level adjust internal trigger DC Level Adjust, is used to set the 
divider output to zero volts, when the CRT trace is 
centered. A small capacitor, C^, provides a first- 
order higher frequency compensation of the divider. 
The signal amplitude is reduced by the divider to 
about one third of the equivalent AC amplitude, which 
reduces the DC mode sensitivity. 



20 



vol tage 
changes 



high- 
frequency 
(HF) reject 
mode 



The DC mode of operation is affected by any change 
in the DC level at the divider output. This could be 
resistance changes due to temperature variations, 
vertical amplifier level changes caused by changing 
"plug-ins" or vertical amplifier drift because of 
aged components. Voltage changes are minimized by 
using precision resistors with low temperature- 
coefficients and by using compensating devices in 
the pickoff circuits, like regulators or pentode 
amplifiers, where small changes in plate-supply 
voltages cause little change in plate current and 
output voltage. 

Instruments that have very high frequency triggering 
capabilities may have a high frequency-reject position 
on the trigger mode switch. This makes it possible 
to reject high frequencies in the presence of low 
frequency signals for more stable triggering at the 
lower frequency. It is particularly useful in areas 
where fast transients are present. A typical trigger 
response is shown by Fig. 2-15. 



10Hz 100Hz 1kHz 10kHz 100kHz 



Fig. 2-15. High frequency reject -trigger 
response. 



function of 
a trigger 
ci rcui t 



When the instrument is switched to HF reject, a 
band width-reduction circuit is added to attenuate 
frequencies above about 100 kHz. A capacitor may be 
added almost any place in the circuit to reduce 
bandpass. One circuit is shown in Fig. 2-16. 

The function of the trigger circuit is to generate 
an output pulse whenever the input voltage level 
reaches a desired level and is changing in the desired 
direction. To do this a level detector, a slope 
detector and a pulse generator are used. 



21 



EXTERNAL 
INPUT 



TRIGGER 
TAKEOFF 



f 



i-fTf 



21 pF 



TRIGGER 
C I RCU I T 



Fig. 2-16. High frequency reject circuit. 

The trigger circuit of an oscilloscope generates 
pulses which activate the sweep generator. The output 
of the sweep generator is then amplified and applied 
to the horizontal deflection plates to produce a 
visible trace on the face of the CRT. The start of 
a display must coincide with the same point on the 
vertical deflection signal with each successive sweep 
if the waveform under observation is to appear stable 
on the CRT face. 



Two general modes of operation are employed to 
establish the coincidence of horizontal and vertical 
deflection signals: 



1. 
2. 



Synchronized 
Triggered 



synchronized In the synchronized mode, the sweep generator is made 
mode to free-run at a frequency just below that of the 

input frequency (or one of its sub-multiples) . The 
input signal, applied directly to the sweep generator, 
then acts to accelerate the free-running condition 
bringing the two frequencies into synchronization. 
A control that changes the sweep hold-off time is 
adjusted to synchronize the sweep. 

triggered Most modern oscilloscopes are operated whenever 
mode possible in the triggered mode. In this mode the 

input signal itself (or some other time related signal) 
initiates each horizontal sweep, thus assuring 
coincidence of the horizontal and vertical deflection 
signals. The sweep generator operates as a monostable 
multivibrator and switches state only upon receipt 
of an external pulse. Once triggered it completes 
its cycle and returns to the original state 
disregarding any triggers, which may arrive during 
the cycling interval. 



22 



The input signal may have a wide variety of shapes 
and amplitudes, many of which are unsuitable as 
sweep initiating triggers. For this reason these 
signals are first applied to a trigger circuit, 
where they are converted to pulses of uniform 
amplitude and shape. 

A typical oscilloscope trigger circuit consists of 
(1) an input amplifier and (2) a pulse generator 
(Fig. 2-17). 



TRIGGER 
SIGNAL 



INPUT 
AMPLIFIER 



PULSE 
GENERATOR 



_»_ TO SWEEP 
GENERATOR 



Fig. 2-17. Pulse generator and trigger circuit. 



input 
amp I if ier 



The input amplifier provides a means by which the 
oscilloscope operator is able to select almost any 
point of a changing slope on the input waveform as 
the starting point for the horizontal sweep. 



23 



INPUT TRIGGERING SIGNALS 



inputs Four sources of triggering signals are usually 

available at the trigger selector switch (Fig. 3-1): 
(1) INTERNAL (INT) (vertical amplifier) (2) INTERNAL 
(plug-in Channel 1) (3) LINE (line voltage, 60 cycles) 
and (4) EXTERNAL (EXT) (external signal required) . 



FREAMPLIFIER 



6.3V, 




VERTICAL 
AMPLIFIER 



TRIGGER 
TAKEOFF 



INTERNAL 

LINE _ 



EXTERNAL Q 



f 




INPUT 
AMPLIFIER 



PULSE 
GENERATOR 



CHANNEL I 
TRIGGER 



Fig. 3-1. Trigger circuit inputs. 



24 



The "INT NORMAL" or Plug-In Position is most often 
used. In this position, a replica of the vertical- 
input signal is applied to the trigger circuit. The 
sweep generator is thus related to the vertical 
waveform under observation. 

Since many signals occur at power line frequencies, 
it is sometimes convenient to trigger the sweep 
"LINE" generator at this rate. In the "LINE" position, 

a replica of the power line voltage, usually from a 
filament winding, is used as a triggering source. 
The 60 Hz waveform is an easy waveform to trigger on 
and by varying the trigger level control, it is 
possible for the operator to vary the relative phase 
of the signal being viewed and its position on the 
face of the CRT. 

Occasions arise when the vertical signal must be 
observed in its relationship to another event, 
possibly occurring at a different time or frequency. 
"EXT" This comparison can be made by using the "EXT" trigger 
position, which connects the trigger input amplifier 
to the "EXT TRIGGER" jack on the front panel. (A high 
input impedance reduces the input amplifier loading 
of the signal source.) 

circuit The Fig. 3-2 is a schematic of a simplified-input 
operation amplifier. This configuration is called a cathode- 
coupled DC amplifier, or voltage comparator. It has 
a high input impedance and presents a minimum load to 
the vertical amplifier or any circuit connected to 
the external trigger input. A circuit of this type 
may be considered a cathode follower driving a 
grounded grid stage, which permits the two cathodes 
to share the same resistor. If both tubes have the 
same transconductance, about one-half of the input 
signal will appear at the cathodes. The common mode 
signal appearing at the cathode subtracts from the 
input signal and reduces the gain of the amplifier. 
This characteristic results in a circuit gain of about 
half the expected value. 



25 



HOOV 



+ioov 



I NPUT O 




OUTPUT 



Fig. 3-2. Simplified- input amplifier. 



The output of this amplifier is limited to two 
extreme conditions: 



the 

extreme 
cond itions 



1. Positive signals of sufficient amplitude 
cause VI to drive V2 into cutoff. The output 
voltage then equals V2's plate supply voltage. 

2. Negative signals of sufficient amplitude 
drive VI into cutoff, and all cathode current 
passes through V2. The output voltage is 
determined by the amount of V2's plate 
current and the resistance of the plate load. 

There is no phase reversal between the input and 
output signal. The output varies over a set range 
and cannot be driven beyond it. The amplitude of a 
useful input signal is limited by the cutoff points 
of VI and V2. When the grids of VI and V2 are at the 
same voltage, the output of the circuit falls halfway 
between the two output limits (in this case 75 V). 



26 



In Fig. 3-3, a variable voltage, ranging from +10 V 
to -10 V is added to the grid of V2. This range is 
sufficient to cut off either VI or V2. Since the 
amplifier is "long-tailed" to the negative 150 V 
supply, the current through the amplifier will remain 
almost constant, regardless of the setting of Rl. 
The output voltage of V2 however, can vary between 
50 V (VI cutoff) and 100 V (V2 cutoff). A 75 volt 
output occurs when both grids are at the same level, 
see Fig. 3-4. Therefore, whenever the input signal 
reaches the same level set at the grid of V2, the 
output from the plate of V2 will pass through the 75 V 
level. In this manner the input amplifier selects a 
point on an input waveform and amplifies it to a 
pre-determined level (provided the amplitude of the 
input voltage does not exceed the range of voltage 
detector available at Rl) . This provides the level detector 
action action necessary in a trigger circuit. 

If it is assumed, as is true in many instruments, that 
the pulse generator will produce a sweep starting 
pulse, only when the output of V2 crosses the 75 V 
level in a negative direction (the actual voltage 
depends on the instrument) ; it follows that only the 
negative-going portion of the input waveform will 
cause this circuit to trigger a sweep, since the input 
amplifier does not invert the phase of the input 
signal. This provides the slope detector action 
necessary in the trigger circuit. 



INPUT 



27 



+ 100V 




>-$ R1 



-150V 



Fig. 3-3. Variable voltage supply. 




INPUT TO VI GRID 



XL 



50V- 



OUTPUT AT V2 PLATE 
Rl SET TO -10V 



100V- 



75V- 



z 



50V- 



7 



OUTPUT AT V2 PLATE 
Rl SET TO O.OV 



100V- 



75V- 



TXJ. 



OUTPUT AT V2 PLATE 
Rl SET TO +10V 



Fig. 3-4. Variable -voltage outputs. 



28 



Reversing the input connections to the grids of VI 
and V2, see Fig. 3-5, reverses the phase of the 
triggering output. The voltage-setting control (usually called 
level the Triggering Level Control) is now connected to 
control the g^d f vi. The pulse generator can then be 
triggered by a positive-going waveform, since the 
input signal phase will be reversed in the plate 
circuit of V2. 

To permit a choice of positive or negative 
"slope" triggering-signals, a "slope" switch is added to the 
switch circuit (Fig. 3-5). This switch couples the input 

signal to either grid, to produce a negative output 

at the plate of V2. 

The Fig. 3-6 incorporates additional circuit elements. 
Resistor R3 limits grid current, when large-amplitude, 
positive input signals are applied in the external 
mode. A small capacitor in parallel with R3 prevents 
attenuation of high frequency signals. 

To provide a constant amplitude output, the input 
amplifier should present the same impedance to 
signals of either polarity. However, high frequency 
"Miller signals are attenuated, when applied to the grid of 
effect" v2 (positive slope position) due to "Miller effect". 
This is feedback, coupled from plate to grid through 
the inter-electrode capacitance of the tube. 

Signals applied to VI' s grid do not experience this 
effect, since its plate is tied to AC ground and 
signals are coupled to V2 through the common cathode. 
However, because V2's plate must be allowed to swing 
if the circuit is to produce an output signal, 
signals applied to V2's grid suffer increasing 
attenuation at high frequencies through plate to grid 
feedback. 



29 



+ 100V 



>. OUTPUT 



I NPUTO- 




TRIGGERING 
>-£ LEVEL 
CONTROL 



Fig. 3-5. Triggering level circuit association. 



+ 100V 

11 



INPUT O CX-O- 



.01 
-* — w\ •- 



-A/W- 

R3 




100V 

A 



>5k 




->- OUTPUT 



: 15k 



-150V 



Fig. 3-6. Slope-switch circuit. 



30 



In Fig. 3-7 the circuit has been balanced by the 
insertion of Resistor R3 and capacitor C2 in the 
plate circuit of VI. C2 approximates the input 
capacity of the following stage. Both tubes now 
present the same impedance to incoming signals and 
the triggering characteristics of the input do not 
change when switching slope. 

Resistors Rl and R2 prevent parasitic oscillations 
'ringing" of the input amplifier and also dampen out any 
"ringing" in the long input leads. 

Capacitor CI improves high frequency gain by 
holding the grid of the control tube at AC ground 
and prevents it from following the cathode signal. 



31 



INPUT O 




OUTPUT 



TRIGGER 
LEVEL 



Fig. 3-7. Trigger circuit for voltage output. 



32 



Improved amplification of the triggering signal is 
achieved in Fig. 3-8. Here the vertical amplifier 
signal is applied in push-pull. Acting as a 
differential amplifier, the input amplifier rejects 
common-mode signals (hum, transients and noise common 
to the grids of VI + V2) and produces a "cleaner" 
signal for the pulse generator. 

Only the AC coupling is shown connected in push-pull, 
however, some oscilloscopes provide the same type of 
coupling in the DC position. 

Small peaking coils and low-value load resistors on 
the plate circuits of VI and V2 extend the high 
frequency response of this input amplifier. 

transistor Transistorized input amplifiers require special 

i nput techniques to overcome the problem presented by their 

amplifier low input impedance characteristic. 



33 



TO VERTICAL _<- 
AMPLIFIER 



+10 



TRIGGERING 
LEVEL 



>1M 



-o*-o — II 



AC 
DC 



LINE 
EXTERNAL 




VI 



TO VERTICAL ,* , 

AMPLIFIER 



.005 



>15k 



-150V 



-CX— O 



AC 
-O DC 



•1M 




V2 



-O LINE 



-O EXTERNAL 



•220 



150 > B3.25uH 



->- +100V 



150$ g3.25yH 



220 



Fig. 3-8. Improved trigger circuit. 



34 



One solution to the problem is shown in Fig. 3-9. 
Here the triggering signals are first applied to a 
cathode follower, which then drives the transistorized 
input amplifier. A field effect transistor may be 
used in place of the tube for the cathode follower 
function. 

Grid limiting-resistor Rl protects VI against 
"base positive signals of excessive amplitude. "Base 
catching" catching" diodes, Dl and D2, limit the range of 
voltage that may be applied to the base of Ql to 
prevent forward or reverse breakdown. (NOTE: 
Temperature compensating diodes have been eliminated 
for simplicity.) In this configuration, the slope 
switch selects the desired polarity from the output 
of the input amplifier. 

Another impedance matching technique is illustrated 
EXT and in Fig. 3-10. The EXT trigger connection has a high- 
INT trigger value series resistance. The INT trigger signal is 
taken from a low impedance point in the vertical 
amplifier. This method does not provide as high an 
input impedance as the vacuum-tube input, but it is 
satisfactory for most applications. The EXT input 
resistor protects Ql and Q2 against excessive input 
voltages, but since it attenuates signals as well, it 
reduces the sensitivity of the triggering circuit in 
the EXT mode. 



35 



+ 12V 



+75V 



INPUTO 




OUTPUT 



Fig. 3-9. Cathode follower input to 
amplifier. 



* Wv i — O* 

1 



EXTERNAL 

WV » Or*— O 

91k 



INTERNAL 



+45V 




It 

I 



47 rr ?47k 



Q1 Q2 




>680 



580pH 



-10V -10V 



■ TRIGGER 



-*- OUTPUT 



Fig. 3-10. Specialized input to amplifier. 



|trig i ist | 




(n* > 


TRIGGER 

SOURCE 

AND 

COUPL 1 NG 

SELECTION 


l 




TRIGGER SIGNAL * 
CHI 


TRIGGER SIGNAL 



Vdi 



TRIGGERED 6-»-0 



1 AUTO \ - 



I 




R3 
200k | 6 . 2k 



Fig. 3-11. Input with operational amplifier. 



37 



input- In some oscilloscopes the input amplifier is classed 
operational as an operational amplifier (Fig. 3-11). 
ampl i f ier 

Diodes Dl and D2 limit the amplitude of the EXT input 
signal to protect transistor Ql. Zener diode D3 
drops the output DC voltage level at the collector 
of Q2 to zero. A portion of the output pulse is fed 
back to the base of Ql in opposite phase to the input 
signal. Diodes D4, D5, D6 and D7 in conjunction with 
their associated components constitute a feedback 
limiter. Action of the circuit is such that the 
output is held to 4 1 V maximum for inputs up to *10 V. 

An advantage of this type of input amplifier lies in 
its stable gain characteristic, which is independent 
of change in transistor gain (Beta) . 



00 



TRIGGER SIGNAL 
CHI 



TRIGGER SIGNAL 




Fig. 3-12. Automatic operational amplifier. 



39 



An additional circuit, shown in Fig. 3-12, provides 
the oscilloscope with an automatic mode of operation. 

When SW1 is placed in the AUTO position, feedback 
automatic resistor R3 is replaced by transistor Q3 and a 180° 
control led phase shift network. The operational amplifier 
amplifier becomes a phase shift oscillator, whose output 

amplitude is still controlled by the limiter circuit. 
The oscillator will free run until an input signal 
of higher frequency is applied to the circuit. Such 
signals will not undergo sufficient phase shift to 
provide positive feedback and will suppress the 
action of the oscillator. The circuit will once again 
function as an operational amplifier and the input 
signal will control the frequency of the pulse 
generator. 



10V 



TRIGGER £20k 
LEVEL 



-10V 



100k 



o wv— 

INPUT 100k 



I 



300k 



10V 



TRIGGER 

LEVEL 

CENTERING 



*- OUTPUT 




■s- 
o 



-10V 



Fig. 3-13. Functional diagram of amplifier 
and comparator. 



41 



Another version of the transistorized input amplifier 
utilizes an operation amplifier and an adder circuit 
(Fig. 3-13). The amplifier provides unity gain for 
leve l the input signal and the trigger LEVEL control signal, 
while the trigger level-centering control signal is 
reduced by 2/3. The sum of these inputs is sent to 
the input amplifier/comparator and the level is 
compared against the ground reference established at 
the undriven base. 



42 



OUTPUT 




Fig. 4-1. Schmitt multivibrator. 



PLATE V2 



CATHODE V1/V2 



GRID V2 



I I 1 



PLATE VI 



GRID VI 




Fig. 4-2. Waveforms of Schmitt multivibrator. 



43 



% 



PULSE GENERATORS 



The purpose of the pulse generator is to provide a 
pulse to the sweep generator each time the signal 
from the input amplifier crosses a specific voltage 
level in a direction as selected by the slope, 
selector. It should operate over a wide range of 
environmental conditions, remain stable as components 
age, regenerate the trigger quickly and respond to 
all frequencies within the vertical bandpass of the 
oscilloscope. 

Most low frequency oscilloscopes (and some early, 
high-frequency models) employ a Schmitt multivibrator 
Schmitt aa a p U i se generator (Fig. 4-1). The "Schmitt" is 
multivibrator a bistable multivibrator and operates as follows: 

Assume VI is conducting and V2 cut off. As the 
grid of VI goes negative in response to the 
input signal, the common cathode follows. At 
the same time, the positive going signal at VI' s 
plate is coupled to the grid of V2. This 
combination of grid and cathode signals drives 
V2 into full conduction, producing a negative 
pulse on its plate. As the input signal returns 
to its reference level (goes positive) VI starts 
to conduct again. The negative signal on its 
plate is coupled to the grid of V2, driving it 
sharply into cutoff. The circuit is then ready 
to accept the next input signal. See Fig. 4-2. 



44 




Fig. 4-3. Hysteresis effect. 



:r7 



o — 

INPUT 
i 




1 

u 



R2? :±C2 



C3 



1 




>R8 
flk 



OUTPUT 




Rl 



R4 ^ / < R5 

TRIGGER 

SENSITIVITY 



Fig. 4-4. Hysteresis adjustment with Schmitt 
multivibrator. 



45 



hysteresis In order to produce a complete cycle (transition) 
VI 's grid must first go sufficiently negative to 
start the regenerative action described above, then 
return to a sufficiently positive level to "flip" the 
multivibrator back to its original state. The voltage 
difference between these two switching levels is a 
measure of the multivibrator's "hysteresis". To 
permit control by low amplitude input signals, a 
multivibrator must therefore exhibit a low hysteresis 
characteristic. See Fig. 4-3. 

The Fig. 4-4 shows a Schmitt multivibrator that 
hysteresis permits adjustment of the circuit's hysteresis. 
adjustment Trigger sensitivity control Rl, inserted between the 
cathodes of VI and V2 now causes the cathode of the 
non-conducting tube to be at a slightly lower potential 
than that of the conducting tube and therefore biased 
closer to conduction. However, the circuit will 
become unstable if the hysteresis is reduced too far. 

Capacitor CI maintains the voltage drop across Rl for 
an instant during transitions, keeping circuit-gain 
high and preserving the fast switching characteristics 
of the multivibrator. 

Since V2 switches between cutoff and a specific 
conducting level, the output pulse maintains the 
same amplitude and polarity, regardless of the 
amplitude of the input signal, (provided the 
period of the input signal remains greater than 
the multivibrator's transition time). 

The output pulse is processed by a differentiator, 
(R8 and C3) so that only the fast rising and falling 
portions of the waveform are passed to the sweep 
generator. 



46 



+,00V +225V 



TRIGGER 
lOOk^" LEVEL 

CENTERING 




OUTPUT 



50V 



Fig. 4-5. Input amplifier. 




Fig. 4-6. Pulse generator. 



47 



At frequencies in the 1-3 MHz range the output pulse 
begins to distort, because the period of the input 
signal becomes less than the multivibrator cycle 
time and the efficiency of the Schmitt multivibrator 
as a pulse generator rapidly deteriorates. Therefore, 
oscilloscopes with this type of pulse generator are 
usually operated in the synchronized mode at 
frequencies above the 5-10 MHz range. A position 
is provided at the trigger mode selector switch 
(labeled "HF SYNC") which capacitively couples the 
input signal directly to the sweep generator. In 
this mode the sweep generator is made to synchronize 
with the input signal. 

NOTE: The sweep generator is made to "lock in" or 
synchronize by adjusting the stability control. 

trigger Stable triggering by low amplitude input signals 
level depends on centering the output of the input amplifier 
centering about the pulse generator's hysteresis range, 
(switching levels). 

To compensate for any mismatches between input 
amplifier and pulse generator in low frequency 
instruments, it is usually only necessary to increase 
the gain of the input amplifier to assure that its 
output crosses the hysteresis range of the pulse 
generator. This method is unsatisfactory in high 
frequency instruments, because it is difficult to get 
both wideband width and high gain. Because of this, 
a Trigger Level Centering adjustment is provided, 
either in the input amplifier (Fig. 4-5) or in the 
pulse generator (Fig. 4-6), to match the operating 
points of the two circuits. 



.p- 

00 



VERTICAL 
AMPLIFIER 




CRT 



TRIGGER 
PICKOFF 



CI 



INPUT 
AMPLIFIER 



HO 



TRIGGERING 
LEVEL 




+225V 



+225V 



C2 

.01 



10k 




VI 



V2 



->- OUTPUT 




♦ — "■ 



+ 1V 



15k 
10mA 



-150V 



2.7M 
— wv- 



+ 125V 



1M< t- 

-12.5V 



;1M 



■150V 



Fig. 4-7. Automatic trigger selection. 



49 



AUTOMATIC If trigger circuit adjustments are not made properly 
OPERATION (and some of them are critical at high frequencies 

and low input amplitudes) no sweep will be generated. 
At the same time it may be necessary to see the input 
signal to determine what adjustments are necessary to 
achieve satisfactory triggering. To provide a sweep 
under these circumstances, two methods are commonly 
employed: The first method places the sweep generator 
in a free-running condition (this method is described 
in the volume on Sweep Generators) . The second method 
involves the trigger circuit. 

A position labeled AUTOMATIC is provided at the 
automatic TRIGGER MODE switch. When placed in this position, 
trigger the switch performs the following operations 
mode (Fig. 4-7) . 

1. Capacitively couples the output of the 
vertical amplifier to the input amplifier 
through CI. 

2. Capacitively couples the input amplifier 
to the pulse generator. 

3. Connects the grids of VI and V2 through a 
resistor. 

4. Sets Trigger Level voltage at zero, (except 
in tunnel diode-pulse generators, not shown). 

NOTE: This arrangement bypasses any front panel 
adjustment that might be improperly set and insures 
a trace regardless of whether or not a signal is 
present. 



Under no-signal conditions the pulse generator free- 
runs at a frequency determined primarily by the time 
constant of the input capacitor and added resistor. 
However, when a signal is present at the input to the 
oscilloscope it will feed through the capacitive 
coupling from the vertical amplifier and input 
amplifier. If its frequency is higher than the free- 
run frequency and has sufficient amplitude, it will 
control the pulse generator operation. 



50 



transistor The Schmitt multivibrator also appears in 

Schmitt transistorized form (Fig. 4-8). Its action is the 

multi- same as the tube version, except when operating in 

vibrator the Automatic mode. 

When the circuit is switched to automatic operation 
by the Mode switch, Q2 conducts, its collector drops 
abruptly about 5 volts and then drops more slowly 
as CI charges. See Fig. 4-9. The base of Q2 goes 
negative until Q2 cuts off, causing an abrupt 5 V 
rise and a slower rise in its collector voltage, as 
CI discharges. As the base of Q2 rises to the point 
of conduction, Q2 turns on again, completing the cycle. 
An input signal to Ql will cause the multivibrator to 
switch prematurely; and as the signal amplitude 
increases it will override the free-running action. 
The multivibrator will then trigger in the normal 
manner, provided the input frequency is greater than 
the automatic free-run frequency. 



51 



o — 

INPUT 
i 



HOOV 



+100V +225V 



C1 
47uF 



+ 100V 
ii 



330 <4.3k 




Ql 




4.3k< -r-Z7 



T 

u i 4 



•1.5k 



i 




OUTPUT 



r 



Q2 



Fig. 4-8. Transistorized Schmitt multivibrator. 




02 EMITTER 



Ql COLLECTOR 




Q2 BASE 



/ 



02 COLLECTOR 



Fig. 4-9. Cycle of Schmitt multivibrator. 



52 



FROM U, 

INPUT < 62k 

AMPLIFIER 

o KM— {>— r 

01 D2 




+ 101 

4-98 



i 22k +10. 
+ 100 



50--> TRIGGER 

[sensitivity 

+ 100V 



f 



Fig. 4-10. High frequency Schraitt trigger. 




BASE OF 
03 



INPUT 
SIGNAL 




IGH-FREqUENCY COUNTOOWN ACTION 



LOW-FREQUENCY ACTION 



Fig. 4-11. High frequency countdown action 
and low frequency action. 



53 



modified In its transistorized version, the Schmitt 
transistor multivibrator can be modified to provide stable 
Schmitt triggering up to and beyond 30 MHz (Fig. 4-10). 
multivibrator Additional circuitry causes it to ignore input pulses 
that arrive during its transition time (a process 
known as "countdown") , so that it produces one output 
pulse for every second, third, fourth, etc., input 
pulse. Its output frequency under these conditions 
is some submultiple of the input signal frequency. 

Circuit operation is as follows: In the quiescent 
state diode Dl is forward biased, D2 is cut off and 
transistor Ql (an operational amplifier) is biased 
below the conducting level. When the positive pulse 
from the input amplifier is applied, Dl disconnects 
and the junction between the diodes goes sufficiently 
positive to turn on D2. The increased drop across 
the base circuit turns on Ql, and the resulting 
negative collector signal is applied to the base of 
Q2, the first stage of the multivibrator. At high 
frequencies, this circuit then begins to count down 
to one of the succeeding pulses and its frequency 
becomes equal to some submultiple of the input 
frequency. See Fig. 4-11. Countdown begins at about 

8 MHz. At these frequencies, Q2 collector load 
increases as inductance (LI) increases. The signal 
voltage swing at the base of Q3 increases to about 

9 volts. The increased signal at the base of Q3 
overdrives the transistor far beyond cutoff in the 
negative excursion and in the positive direction 
lifts Q3 emitter until Q2 is cutoff. Under these 
conditions the multivibrator cannot be driven until 
the LR time of LR1 returns the base of Q3 to its 
normal operating level. D3 and D4 are reverse 
breakdown protection diodes for Q2 and Q3. 

However, since a large number of triggers are still 
generated in the interval occupied by even the 
fastest available sweep, countdown does not result 
in a loss of repetition rate. 



54 



+ioov 
1 



R2 

i an 



25k 



.001 



TRIGGER ""? " ®) 
SENSITIVITY 



■ 68k 




X 



D4 T ' 5pF OUTPUT 



Fig. 4-12. Tunnel diode pulse generator. 



FORWARD CURRENT 
IN MILL I AMPERES 



AC LOAD LINE 
COMPOSED OF 
L1, R2, R3, R4 Cj 




FORWARD VOLTAGE 
IN MILLIVOLTS 



Fig. 4-13. Tunnel diode characteristics. 



55 



tunnel diode The tunnel diode (TD) pulse generator permits 

pu I se triggered oscilloscope operations at extended high 

generator frequencies. It is characterized by its simple 

circuitry and its ability to operate on very fast 
low energy-input pulses. The TD pulse generator 
(Fig. 4-12) is used extensively in wide band 
oscilloscopes and permits triggered operation over 
the entire bandpass of the instrument. The tunnel 
diode pulse generator is preceded by an input 
amplifier using high transconductance (g m ) tubes to 
provide sufficient switching current for this mode. 
(Since it came into use before the modified 
transistorized Schmitt, the TD pulse generator may 
also be found in some relatively low frequency 
instruments. ) 

The Trigger Sensitivity control is set so that in 
the quiescent state about 9 mA of current passes 
through tunnel diode D3. The diode rests below point 
A on the first slope of the tunnel diode curve in 
Fig. 4-13. The input terminal is slightly positive 
and about 2 mA of current passes through diode Dl. 
As the negative signal from the input amplifier cuts 
off Dl, an additional 2 mA of current passes through 
D3 switching it to its high state, point C in 
Fig. 4-13. As the current through LI increases, the 
current through D3 shifts down the tunnel diode curve 
to point B and rests conducting about 1 mA of current. 
When the input signal goes positive, Dl again 
conducts, the current through D3 is reduced, and D3 
drops back to its low voltage state. The impedance 
of D3 in the low state is about 8 ohms, therefore 
the current increases to 9 mA, placing the operating 
point just below point A which completes the cycle. 
As the input frequency increases, inductor LI causes 
current in R2 and R3 to lag, and the circuit begins 
to countdown. (Countdown usually occurs between 2 
and 6 MHz, but higher input amplitudes may cause it 
to take place at higher frequencies.) The voltage 
change across the tunnel diode is about half a volt 
and may require amplification before being applied to 
the sweep generator circuit. Transistor Ql performs 
this function, accepting the TD output and applying 
amplified pulses to the sweep generator circuit 
through transformer Tl. Diode D4 is placed across 
Tl's secondary winding to limit the output pulses to 
a single polarity and prevent ringing. 



56 



+225 

+ 100 \jf TRIGGER 

SENSITIVITY 



+225 



NPUT |R225^| ™^ R £ fj iR gk 




Ql 



AMPLIFIER J 200 f CENTERING 
1k 



+ 100 



01 @ 




Q2 



470> J12k 



HI - 



TO SWEEP 
GENERATOR 



T1 



Fig. 4-14. Tunnel diode pulse generator 

circuit with transistor driver. 



57 



The Fig. 4-14 illustrates another type of TD pulse 
generator which eliminates the need for high 
transconductance tubes in the input amplifier. 
Transistors Ql and Q2 form a current amplifier, 
converting the push-pull output of the input 
amplifier to a current driving signal for tunnel 
diode Dl. Maximum current is limited by long-tailed 
emitter resistor Rl. The series inductance of LI 
and Tl, togehter with the output resistance, control 
the countdown frequency of the circuit. 

NOTE: The second output from this pulse generator 
from Tl controls the action of a multivibrator in the 
associated sweep generator. In the absence of an 
output from transformer Tl (automatic mode) , the 
sweep generator free runs, providing an untriggered 
sweep for the CRT. (For complete details, see volume 
on Sweep Generators.) 



58 



TRIGGER 
S I GNAL 



TRIGGER 




DELAYING 

SWEEP 
GENERATOR 


1 DELAYING SWEEP 

< DELAY TIME 
*■< MULTIPLIER 

| ENABLING PULSE 






















DELAY 
P 1 CKOFF 








1 








' 


/\ DELAYED SWEEP 




TRIGGER 




DELAYED 

SWEEP 

GENERATOR 


HORIZONTAL 
AMPLIFIER 




-44- 
















VLJV 



Fig. 5-1. Schematic for delayed sweep operation. 



59 



DELAYING AND DELAYED SWEEPS 



delay Instruments with delayed sweep capabilities employ 
pickoff a special trigger circuit called a "Delay Pickoff" 
circuits (Fig. 5-1). Its purpose is to start or enable a 
second (delayed) sweep at a precisely controlled 
time, measured from the start of the first (delaying) 
sweep . 

The following section on delaying and delayed sweeps 
is developed around the triggering and gating circuits 
involved. The discussion of circuits used to develop 
the actual sweep waveform (sawtooth) is reserved for 
the volume titled, "Oscilloscope Sweep Circuits." 

Delaying-sweep measurements are based on the use of 
delaying and two l inear calibrated sweeps. The first sweep, 
delayed commonly called the delaying sweep, allows the 
sweeps operator to select a specific delay time. When this 
time is reached, the delayed sweep starts. The 
delayed sweep on some instruments is faster than 
the delaying sweep. It offers additional resolution 
and increases the accuracy of time-interval 
measurement. 



60 



To understand delaying sweep operation, it is 
necessary to understand the time relationship between 
the delaying sweep and the delayed sweep. To 
illustrate, an event occurs, that starts the delaying 
sweep at point tg. The delaying-sweep voltage ramp 
is applied to a voltage comparator, that produces 
an enabling pulse later in time, t^ . This pulse 
occurring at tj starts the delayed sweep. Then the 
delay time may be defined as the difference in time 
between the start of the delaying sweep and the 
start of the delayed sweep and can be expressed as 
ti - tg. See Fig. 5-1, 5-2 and 5-3. 

Before discussing how a delaying or delayed sweep is 
generated, it would be wise to discuss applications 
where they are useful. 

There are three basic types of measurement that can 

be made with a delaying sweep, arranged by order of 

increasing accuracy, they are: (A) absolute delay, 

(B) incremental delay, and (C) ratio measurements. 

(A) Absolute delay as shown in Fig. 5-4A is the 
measurement of time from the start of the delaying 
sweep to some point along the delaying sweep. 

(B) Incremental delay is the measurement of two 
absolute delays and finding the difference between 
them. Fig. 5-4B. 



1 „ 




DELAY TIME 
T T 3 




, 1 




1 1 
1 1 
1 1 
1 1 




r*- 




H 




jL 


DELAY 1 NG 

SWEEP 


_U-"TL 


DELAY 

PICKOFF 

VOLTAGE 

COMPARATOR 


J. 


DELAYED 
SWEEP 










i 
i 

i i / 

i i 
i i > 


INTENSIFYING 




' 


r 




1 




\ 


PULSE 






HORIZONTAL 
AMPLIFIER 






TRACE 
INTENS IFIER 


CRT 
























, 


, 





Fig. 5-2. Schematic for delaying sweep 
operation. 



61 



UNBLANKING 
COMPOS I TE 
WAVE SHAPE 




DELAYED 

SWEEP 



Fig. 5-3. Delaying sweep oscilloscope circuit 
relationships . 



n 



-ABSOLUTE DELAY *-j 

START OF SWEEP 



-DELAY 1- 



• DELAY 2 ■ 



START OF SWEEP 



I PULSE WIDTH 
START OF SWEEP 



INCREMENTAL DELAY 
DELAY 2 - DELAY 1 " 



RATIO = 



PULSE WIDTH 2- 



PULSE WIDTH 1 
PULSE WIDTH 2 



(A) 



(B) 



(C) 



Fig. 5-4. The basic types of measurement using 
delaying sweep techniques. 



62 



(C) A ratio measurement takes the ratio between 
two incremental delays. Fig. 5-4C. 

de I ayed An absolute delay measurement may be made between 
trigger two independent signals, one which externally triggers 
the sweep and one occurring later in time which is 
displayed. This mode may also be used to generate 
a delayed pulse to trigger another piece of equipment 
from the DELAYED TRIGGER output of the CRO. The 
accuracy of this type of measurement is limited by 
the processing time of the trigger signal, the 
slope of the trigger signal, the processing time of 
the pickoff circuit and the rate and linearity errors 
in the delaying sweep. 

Except for the rate and linearity errors in the 
delaying sweep, the above errors are not particularly 
significant until one begins to use the faster sweep 
speeds (i.e. small absolute delay measurements). For 
most situations, this method of measurement of 
absolute delay is more accurate than making the 
measurement from the graticule. 

delay time- The delaying sweep, Delay Time-Multiplier (DTM) 
multiplier potentiometer used to measure delay times, also 
potentiometer offers greater resolution under most conditions, 

than just reading the delay times from the graticule. 

This improved resolution also applies when generating 

a delayed pulse as this allows more precise time 

interval settings. 

A measurement made using the CRT graticule depends 
upon the oscilloscope's ability to line-up precision 
time-markers with major graticule divisions, and is 
subject to errors contributed by non-linearity in 
the horizontal amplifier and CRT. The delaying- 
sweep method uses the CRT as a null read-out device 
that does not affect the measurement accuracy. 

An intensifying pulse applied to the CRT indicates 
when the delayed sweep starts with respect to the 
delaying sweep, so delay time can be determined 
independently of horizontal amplifier and CRT 
considerations. The portion of the delaying sweep 
that is intensified is a direct function of the 
duration of the delayed sweep as shown in Fig. 5-3. 



63 



error of The possible error of the graticule method is 
graticule determined by the accuracy of the sweep as observed 
method on the graticule. With a specification of ±3% of 
full scale, a measurement has a maximum possible 
error of *0.3 ms on the 1 ms/sec range. This type 
of error is easy to calculate because the ±3% 
specification normally includes both a timing or 
rate error and a nonlinear ity factor. The delaying- 
sweep method also has these same basic factors which 
contribute to possible errors, although they are 
more complex in nature. Since the voltage ramp of 
the delaying sweep is the time base used in the 
delaying-sweep method, the basic sources of error in 
time-interval measurements can be illustrated as 
shown in Fig. 5-5. 




DELAY TIME 



Fig. 5-5. Error in time- interval measurements. 



64 



The dotted line represents a no-error condition, 
while the linear solid line represents the average 
rate error due to differences of the timing networks 
nonlinearity in the delaying sweep. The nonlinearity factor varies 
about the solid line and represents the actual delay 
time read on the DTM. This nonlinearity factor is 
a combination of both sweep and potentiometer 
nonlinearity, and is usually expressed in terms of 
dial divisions on the DTM. Delaying-sweep percentage 
error is then calculated by using this figure and 
the percentage figure for rate error. Percentages 
of rate error are typically expressed in terms of 
actual delay time and are usually better than the 
sweep-rate accuracy as read on the CRT, because they 
do not depend on the horizontal amplifier and CRT. 

Absolute delay is a measurement from the start of 
the sweep to the occurrence of an event. 

Since the trigger circuit is a level selector, it 
is difficult to tell exactly where the point T is, 
especially when the triggering signal has an a 
appreciable slope. The delay adjustment cannot be 
reduced to zero to check the start point and the 
accuracy of the delay start and stop adjustments 
affect the accuracy of the delay readings. There 
is also a time delay in the trigger circuit from the 
pickoff point in the vertical, to the starting point 
on the CRT. This delay may change as the triggering 
source is changed from main amplifier to plug-in 
triggering or to external triggering — these delays 
are especially important at higher sweep speeds as 
they may become an appreciable part of the 
measurement . 

Fixed delay is a result of inherent circuit delays. 
fixed delay because the comparator requires time to generate a 
trigger pulse, and the delayed sweep and trace 
intensifier require time to generate an intensifying 
pulse. At these faster sweep rates, the delay time 
can be expressed as accurate within some percentage 
plus some fixed delay such as 100 ns. See Fig. 5-6. 

One source of error mentioned is the delay time 
multiplier and is due to nonlinearities of the 
potentiometer resistance element and the mechanical 
linkages. The comparator itself, due to possible 
changes in firing threshold at various voltage 
levels, may contribute some error. 



65 





NONLINEARITY // 


AVERAGE 




FACTOR v. / / 


RATE 
ERROR 




jr 






y7 s 






SI y 


















/ / / 






/ / y 






/ / / 






/ jS / 






/^ / 






_-^^ / 






Ss s 






r / s 






/ / 






/ / 






s y 




//s' 






f/y 







FIXED 

DELAY 



DELAY TIME 



Fig. 5-6. Error in time- interval measurements 
with fixed delay. 



Incremental delay measurements are made between two 
points along the delaying sweep. The time measurement 
is the difference between two absolute delays. A 
measurement of this type is typically made between 
the leading and trailing edge of a pulse, the leading 
edges of two pulses, or any two points of interest 
along a waveform. 

The example in Fig. 5-7 illustrates a typical 
incremental measurement. Note the intensified portion 
of the lower waveform which is displayed in magnified 






Fig. 5-7. Determining the delay time of 
pulse 1. 



66 



VARIABLE 

TIME/CM o« DELAY TIME f 



DELAY-TIME MULTIPLIER 
1-10 




Fig. 5-8. Delay time reading 

Iras x 3.27 = 327 ms. 




Fig. 5-9. Determining the delay time of 
pulse 2. 




Fig. 5-10. Determining time between events by 
graticule measurement. 



DTM 



67 



form above it. The delay time associated with this 
event is determined by multiplying the delaying 
sweep rate of 1 ms by the DTM reading of 3.27, as 
shown in Fig. 5-8. By turning the DTM until the 
next event is intensified and the upper waveform is 
in the same relative graticule position as before, 
a display like Fig. 5-9 is seen. If the delay time 
of this event is 7.47 ms, then the time interval may 
be computed. 

It is not necessary to determine the actual delay 
time of the start of the event, however, in both 
cases the DTM is adjusted until the magnified 
portion of both events is in the same relative 
graticule position. In the examples shown, the 
center (or 5-cm point) of the graticule is used. 
Any point on the graticule may be used as long as 
it is the same point for both events. 

The difference between these two delay times is 
4.20 ms and corresponds to the period of time between 
the two intensified events. Each minor division on 
the DTM represents 0.01 ms, which also represents the 
resolution of the delay time. Measuring with the 
graticule, the resolution is approximately 0.1 ms or 
reduced by a factor of ten. Fig. 5-10 shows the same 
two events with a sweep rate of 1 ms/cm. 

In Fig. 5-10, each minor horizontal graticule division 
corresponds to 0.2 ms . The time between the two 
events, as read from the graticule, is 4.2 ms. When 
making a graticule measurement, only two numbers are 
significant because of the limitations in resolution 
due to trace width and display size. Thus the 
delaying-sweep method offers the additional 
resolution of an extra significant number. 

By using the incremental measurement technique, the 
uncertainty of the measurement start point T c is 
removed when the first measurement is subtracted from 
the second measurement. 

(T - T 2 ) - (T - T 2 ) = T 2 - T 2 

So T is no longer included in the computation, but 
the processing time of the trigger circuits is a 
constant and does not vary between measurements. 



68 



Error percentages are related to full scale readings 
and therefore become a larger percentage of a small 
measurement. If we say that a sweep accuracy Is ±1% 
this means that there may be an error of ±1 mm in ten 
centimeters — it also infers that there might be a 
±1 mm error any place along the sweep and still be 
within specifications. 



incrementa I 
measurement 
technique 



In making incremental measurements, it is found that 
the error reduces at about the same rate as the 
measurement and the 1% error is 1% of the measurement 
made and not as great as 1% of full scale. The error 
due to helidial nonlinearity is ± 2 minor divisions , 
however, and may contribute an error of 2% for a 1 cm 
or a 1 major division measurement. This is a fixed 
potentiometer linearity specification and becomes a 
smaller part of the measurement as the measurement 
is made over a larger portion of the CRT or delay 
time multiplier dial range. Maximum error may then 
be specified as: 



= *(1 +f) % 

1 The 1% accuracy of the sweep 

2 The ±2 minor divisions of DTM pot linearity 
A The actual difference reading in major 

divisions or DTM revolutions. 

The error is graphically produced in Fig. 5-11. 



MAXIMUM 
ERROR 
AS i OF A z 



vl Mill L' 



A IS THE DIFFERENCE BETWEEN 
TWO DELAY-TIME MULTIPLIER READINGS 



Fig. 5-11. Error when using incremental 
measurement technique. 



69 



From the graph it becomes obvious that greater 
accuracy is obtained by making a measurement over 
as large a part of the graticule or delay time 
control as possible to reduce the error contributed 
by delay time potentiometer nonlinearity. It should 
seldom be necessary to make a measurement less than 
about 3 cm because the greatest next sweep speed 
ratio will expand this to 8 cm by going to the next 
higher sweep speed on the delayed sweep. 

ratio The most accurate delaying sweep measurement that 
measurement can be made is a ratio measurement. 

Ratio measurements are made, when two time periods 

such as two pulse widths are compared. This is 

actually two incremental measurements. In this 

mode, the results of the two incremental measurements 

. Measurement #1 

are compared — ,,„ and the ratio cancels out 

Measurement #2 

the sweep speed times. The result, however, is no 

longer a time measurement, but a ratio between the 

two time periods. The exact time of each occurrence 

is not used, but rather the ratio of one to the other. 

This is quite accurate if measurement is made over a 

large part of the Delay Time Multiplier dial. 

It should be emphasized that all of these are worst 
case errors and will typically be less in actual 
practice. Delaying sweep accuracies are not ±1% on 
all instruments which may make results vary from the 
examples given. Specifications for the instrument 
being used should be checked. 

The resolution of these delay times can be improved, 
thus improving the time-interval measurement accuracy, 
by driving the horizontal amplifier with the delayed 
sweep. The intensified portion of the delaying-sweep 
presentation is now displayed over the full CRT 
display. In the case of Fig. 5-3, this appears as a 
10X magnified display since 1/10 of the original 
waveshape time is now displayed over the same 
graticule area. A delaying-sweep oscilloscope then 
acts as a magnifier, whose magnification power is the 
ratio of the delaying-sweep rate to the delayed-sweep 
rate. 



70 



, ., . delaying sweep rate 

Magnification * ■, j 7ZTZ 

6 delayed sweep rate 

1 ms/cm (Sweep rates used 
1 s/cm are only 
examples) 

10~ 3 
10 -6 

1000:1 

Fig. 5-12 illustrates the use of delaying sweep to 
magnify a signal 1000X to allow closer examination 
of leading edge detail. Some oscilloscopes can 
provide a dual display of these two sweep rates. 
This is accomplished by using an internal 
multivibrator to switch between sweep rates and is 
referred to as automatic display switching. 

When measuring short time intervals, it is not always 
possible to use the delaying sweep at a fast enough 
rate and still maintain a proper delaying sweep to 
delayed sweep ratio for the desired magnification. 
When this occurs, the graticule method may prove 
more accurate. After using the delaying-sweep method, 
simply multiply the measured time interval by the 
delaying-sweep rate accuracy (as in the graticule 
method) and use the most accurate figure. In most 
cases, the delaying-sweep method will be the more 
accurate way of making time- interval measurements. 

improving Accuracy can be further improved by using a time-mark 
accuracy generator to calibrate the DTM. By selecting time- 
marks so that accurate time-marks can be intensified 
and magnified for each ten major dial divisions, a 
calibration chart can be constructed for any delaying 
sweep rate that is needed. 

The accuracy of this final measurement is limited 
only by the accuracy of the time-markers and the 
nonlinearity that occurred in 10 dial revolutions of 
the delay time multiplier dial. This accuracy can 
be held to about ±0.1%. 



71 



ii 



Fig. 5-12. Magnified 1000X display resolves two 
pulses from apparent single pulse. 



72 



DELAYING SWEEP 





INPUT 






PULSE 
GENERATOR 




CF/EF 






AMPLIFIER 














+ 










n 












"*1 


: DELAY-TIME 
I MULTIPLIER 















Fig. 6-1. Delaying sweep generator. 



DELAYING SWEEP 
SAWTOOTH INPUT 



+225V 




Fig. 6-2. Delay time multiplier and input 
amplifier. 



73 



TRIGGERED DELAYED SWEEPS 



In the type of operation just described, the delayed 
sweep is operated in a free running mode and starts 
when it receives an enabling pulse from the delay 
pickoff. If there is time jitter between the 
delaying sweep trigger signal and the signal viewed 
with the delayed sweep , it will be magnified by the 
ratio of the two sweep speeds . 

Sometimes it is useful to look at a signal that has 
time jitter time jitter and to be able to hold it stationary on 
the screen. Operation of the delayed sweep in a 
triggered mode allows this. With this configuration, 
the delaying sweep produces an enabling pulse in the 
normal manner, but the delayed sweep is only armed 
and will not sweep until it receives a trigger pulse 
from the delayed sweep trigger circuit. As a result, 
the delayed sweep starts with the observed pulse and 
is not time related to the pulse that triggered the 
delaying sweep. This causes a jittering pulse to 
appear jitter free. Precision time measurements 
using the delay time multiplier cannot be made in 
this mode because the time delay is only partially 
dependent on the delay time enabling pulse. When 
the display is observed in the B Intensified by A 
position of the Horizontal Display switch, the 
brightened part of the sweep will appear to jump from 
waveshape to waveshape as the Delay Time Multiplier 
dial is rotated. 

The input signal to a Delay Pickoff is the sawtooth 
triggering output voltage supplied by the delaying sweep 
level control generator (Fig. 6-1). Since any point on the ramp 
of the sawtooth may represent the desired delay 
between sweep starts, the triggering level control 
(now called a Delay Time Multiplier) in the input 
amplifier must supply a range of voltage equal to the 
10 major-division value, of the unattenuated sawtooth 
(Fig. 6-2). A multi-turn precision potentiometer is 
used for this purpose and its output is applied to 
the grid of V2. 



74 



Because the input amplifier must respond to such a 
wide range of input voltages, tube V3 is placed in 
the cathode circuit of VI and V2 to provide a source 
of constant current. 

Since the rate of rise of the input sawtooth is 
relatively low, the input amplifier is designed for 
high gain. The resulting high amplitude output 
signal results in more positive switching action in 
the pulse generator. No Trigger Level Center 
delay adjustment may be found in this circuit, but the 
start/stop Delay Start and Delay Stop adjustments perform the 
same function. The Delay Start potentiometer is 
adjusted so that the pulse generator produces an 
output when the input sawtooth has risen to 10% of 
its 10 major division voltage, and the delay time 
multiplier is set at a value of 1.00. The Delay 
Stop potentiometer is set to produce a pulse when 
delay the input sawtooth rises to 90% of its 10 major 
start/stop division value and the Delay Time Multiplier is set 
accuracy at 9.00. (These adjustments interact and require 
several adjustments for accurate delay timing) . 
Each major division on the delay time multiplier now 
represents one-tenth of the delaying sweep time (one 
major division on the graticule) . The accuracy of 
the delay pickoff circuit depends largely on the 
linearity of the delaying sweep sawtooth and the 
Delay Time Multiplier potentiometer. 

input Fig. 6-3 is a schematic of a hybrid input amplifier, 
amplifier in which a transistor replaces the vacuum tube as a 
constant-current source. 

delay Fig. 6-4 represents a fully transistorized delay 
pickoff pickoff circuit. Diodes Dl and D2 protect transistors 
circuit Ql and Q2 against Emitter-Base Junction reverse 

breakdown, when they are not conducting. Tunnel 

diode D3 acts as a pulse generator. 



75 



DELAYING SWEEP 
SAWTOOTH INPUT 




MULTIPLIER 



Fig. 6-3. Transistor/ tube hybrid input 
amplifier. 



DELAYING SWEEP 
SAWTOOTH INPUT 




^S DELAY TIME 
^^ MULTIPLIER 



Fig. 6-4. Transistor delay pickoff circuit. 



76 



A delay pickoff circuit may also utilize a Schmitt 
pulse multivibrator as a pulse generator (Fig. 6-5). Since 
generator the delayed trigger pulse is often made available at 
the front panel for use on external equipment, a 
low impedance output is required to prevent loading 
low of the pulse generator. Cathode follower V3 provides 

impedance this low impedance and, through the action of the 

coupling capacitor and resistors in its grid circuit, 
differentiates the delayed trigger pulse to provide 
a more narrow output pulse. The resistors are so 
zero selected that V3 is biased at cutoff, providing a 
baseline zero baseline voltage for the output pulse. 

In Fig. 6-6 a tunnel diode pulse generator is followed 
by an amplifying transistor. The emitter-follower 
configuration provides a low impedance output and, 
since it is not forward biased, also establishes a 
zero baseline voltage for the output pulse. The 
capacitors and resistors in the base circuit of Q5 
also form a differentiator to shape the output pulse. 



77 



+350V 




1 




+ 100V 



MOk 



95k? -^27 

T /Ct\ 47 P F 




T 

50V 



_ DELAYED TRIGGER 
OUTPUT 



Fig. 6-5. Schmitt trigger delay pickoff circuit. 




INPUT AMPLIFIER/COMPARATOR 



»- OUTPUT 



+ I00V 



Fig. 6-6. Low- impedance output from tunnel 
diode pulse generator. 



78 



If the output of the emitter follower Is fed back to 
monostable the emitter of the amplifier, as in Fig. 6-7, the 
circuit becomes a monostable multivibrator. This 
circuit produces one output pulse of rapid and clear 
definition from one input pulse and operates as 
follows: 

With no input, Q4 is conducting about 0.5 mA 
and Q5 is turned off with a negative 0.5 V 
on its base. A negative pulse from the tunnel 
diode turns Q4 on harder. A 12 V pulse from 
the collector of Q4 turns Q5 on, which feeds 
a 10 V pulse back to the emitter of Q4 and 
saturates Q4. The RC time of the feedback 
network between the emitters keeps Q4 
saturated for about 1 us before it reverts to 
the original state. 



+225V 
i 



79 



; 470 + 100V 



"©- 



rmr~ 



INPUT 
COMPARATOR 



! 180k 



200 




QA 



I 



120 
001 




Fig. 6-7. Monostable multivibrator. 



81 



INDEX 



Absolute delay, 60, 64 
AC coupling, 2, 15, 32 
Amplifier, 

automatically controlled, 39 

bandwidth, 2 

bandpass, 2, 12 

cascode, 11 

gain, 2, 15 

horizontal, 3 

hybrid, 74 

pentode, 20 

transistor, 32, 37, 74, 76 

valve, 73, 74 

vertical, 2 
DC coupling, 2, 15, 19 
Delay line, 9 
Delay pickoff, 59 
Delay start/stop, 74 
Delay time multiplier, 62 
Delayed sweep, 59 
Delaying sweep, 59 
Error, 

gauss ion curve, 2 

graticule measurement, 63 

incremental measurement, 68 

ratio measurement, 70 
Frequency, 

high-frequency reject, 20 

low-frequency reject, 17 

wide frequency response, 9 
Hysteresis, 45 
Hysteresis adjustment, 45 
Inputs , 23 
Measurement, 

graticule, 63-66 

incremental, 66-69 

ratio, 69-70 
Miller effect, 28 
Pulse generators, 43 

tunnel diode, 55-58 
Ringing, 30 



Schmitt multivibrator, 43 

transistor, 50-53 
Slope width, 28 
Sweep generator, 3 
Synchronized mjde, 9, 21 
Time jitter, 73 
Trigger, 

delayed, 62 

external, 11, 34 

internal , 34 

level centering, 47 
Triggering, 

automatic, 49 

DC, 12 

DC from channel 1, 13 

delayed sweep, 73 

level control, 28 

normal, 13 

signals, 23 
Trigger takeoff, 

amplifier, 5 

coupling, 5 

noise, 5 

pulse generator, 5 

single channel, 11 

voltage changes, 20 

Y-T displays, 3 



83 



CIRCUIT CONCEPTS 
BY INSTRUMENT 



Most concepts discussed in this book are applicable 
to all Tektronix products containing a power supply. 
However, the concepts listed on the following pages 
are used in only certain products. Applicability is 
indicated by a • . 



84 



CIRCUIT CONCEPTS: 


INSTRUMENTS: 

OSCILLOSCOPES: 

< <i 

O t^ i-l CM CO ■* i-t 
r-i rH CM CM U-l U-l 0\ 

m m cn <t -3 <t <t 


<! 

CM 
O 

in 


CO 

o 
in 


o 
in 


in 

i-l 
m 




Absolute Delay 
(p 60) 


























Automatic Operation 
(P 49) 


























Delay Pickoff Circuits 
(P 59) 


























Delay Pickoff Circuits, 
Transistorized (p 74) 


























Delayed Sweep Accuracy 
(p 60) 


























Hysteresis Adjustment 
(P 45) 




• 


















• 




Incremental Delay 
(p 60) 


























Input Amplifier 
(p 22) 


























Ratio Measurement 
(p 62) 


























Schmitt Multivibrator 
(P 43) 


• 


• 


• 










• 


• 


• 


• 




Single-Channel Amplifier Bandpass 
(P 12) 


























Single-Channel DC Triggering 
(v 13) 


























Single-Channel Takeoff Circuit 
(p ID 


























Synchronized Operation 
(p 21) 


• 


• 












• 






• 




Trigger Coupling Circuit 
(P 15) 


























Trigger Level Centering 
(P 47) 


• 


• 


















• 




Trigger Takeoff Circuits 
(p 5) 


























Triggered Delayed Sweep 
(P 73) 


























Tunnel Diode Pulse Generator 
(P 55) 








• 



















85 



PLUG- INS : 



< < 
en m vd 
en en en en 



< 
en 
-tf 



pq 



■* -a- 



u-i in 

u~l u~i v£> 00 



mirjiomioirtiniriiriinioiriminio 



u-i 
co 



H N ^ 
„ r < < <! 

S H rH rH rH 



• ••••••• • 

• ••• ••••••••« • 

• ••••••• « 

• ••••••• « 

• ••••••• • 

1 1 

••••(•• •• • 

7T"777T77T - 7 

1 

• ••• ••••••••••• • 

• ••••••• • 

• ••• ••• •• « • 

• • • • 

• • • • 

• •••••• • •• • 

• ••••••••••••••• • 

• • • • ••• •••••• • 

• ••••••••••••••• • ••• 

• ••••••• • 

• • • • • I 



86 



CIRCUIT CONCEPTS: 


INSTRUMEN' 
PLUG-INS continuec 

O H CO 1^ 

\0 ^O v£> vD i— 1 CM 

<J <i <! ca < < 

CM CN CM CM CO CO 


rs: 

i: 

CO 

< 

CO 


3 

CO 


< 

CO 


«: 

CO 


00 

< 

CO 




Absolute Delay 
(p 60) 


























Automatic Operation 
(P 49) 








• 


















Delay Pickoff Circuits 
(d 59) 


























Delay Pickoff Circuits, 
Transistorized (p 74) 


























Delayed Sweep Accuracy 
(p 60) 


























Hysteresis Adjustment 
(o 45) 


























Incremental Delay 
(P 60) 


























Input Amplifier 
(P 22) 








• 


















Ratio Measurement 
(v 62) 


























Schmitt Multivibrator 
(p 43) 








• 


















Single-Channel Amplifier Bandpass 
(P 12) 










• 








• 








Single-Channel DC Triggering 
(p 13) 










• 








• 








Single-Channel Takeoff Circuit 
(P ID 










• 








• 








Synchronized Operation 
(P 21) 


























Trigger Coupling Circuit 
(p 15) 








• 


















Trigger Level Centering 
(p 47) 


























Trigger Takeoff Circuits 
(d 5) 


























Triggered Delayed Sweep 


























Tunnel Diode Pulse Generator 
(P 55) 



























87 





















<$ 




<1 


CNI 


■* 


m 










U2 


rH 


CN 


rH 


CM 


P^ 


r^ 


r^ 


(SI 


co 


<r 


m 


VO 


<! 


<i 


PQ 


co 


< 


< 


<! 


w 


cu 


CO 


M 


c> 


O 


o 


rH 


rH 


m 


co 


co 


co 


CO 


co 


en 


en 


H 


H 


t-l 


rH 



• • • • • • 

1 

• • • • • • 

• • • • • • 

• • • • 

• • • • • • 

• • • • • • 



NOTES 



NOTES 



This book is but one of a series. 
The series consists of two groups, 
circuits and measurements . These 
texts present a conceptual approach 
to circuits and measurements which 
apply to Tektronix products. 
Several "concept books" are in 
preparation; those now available 
are: 



Power Supply Circuits 
062-0888-01 

Oscilloscope Cathode-Ray Tubes 
062-0852-01 

Storage Cathode-Ray Tubes and Circuits 
062-0861-01 

Television Waveform Processing Circuits 
062-0955-00 

Information Display Concepts 
062-1005-00 

Semiconductor Devices 
062-1009-00 

Spectrum Analyzer Circuits 
062-1055-00 

Oscilloscope Trigger Circuits 
062-1056-00